Multi-rate system for audio processing

ABSTRACT

A multi-rate audio processing system and method provides real-time measurement and processing of amplitude/phase changes in the transition band of the lowest frequency subband caused by the audio processing that can be used to apply amplitude/phase compensation to the higher subband(s). Tone signals may be injected into the transition band to provide strong tonal content for measurement and processing. The real-time measurement and compensation adapts to time-varying amplitude/phase changes regardless of the source of the change (e.g. non-linear time-varying linear or user control parameters) and provides universal applicability for any linear audio processing.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation application of U.S. Ser. No.14/824,998 entitled ‘MULTI-RATE SYSTEM FOR AUDIO PROCESSING” filed onAug. 12, 2015, now allowed, which claims benefit of priority under 35U.S.C. 119(e) to U.S. Provisional Application Nos. 62/115,560 entitled“MULTI-RATE SYSTEM FOR ARBITRARY AUDIO PROCESSING” filed on Feb. 12,2015, the entire contents of which are incorporated by reference.

BACKGROUND

Audio reproduction hardware that supports high sampling rate content iscurrently in use. However, audio processing techniques may not berequired to process the full bandwidth of such content. Because audioprocessing at higher sampling rates requires greater computationalresources, it is undesirable to perform audio processing designed forlower sampling rates at higher sampling rates. This is especiallycritical for systems having limited computational resources.

Various audio processing techniques use multi-rate processing because ofits computational advantages. In general, multi-rate subband processingincludes subband decomposition, decimation, and expansion processes.These processes allow signals to be processed at reduced sampling ratescorresponding to the bandwidth of the subbands while preserving the fullbandwidth of the original content in the reconstruction phase.

Some techniques apply audio processing only to the lowest frequencysubband of the subbands in order to reduce computational complexity.However, performing audio processing in the lowest frequency subbandoften causes amplitude and phase changes. Existing techniques attempt tocorrect for these changes by adding compensation filters in the higherfrequency subbands that match amplitude and phase in the transition bandof the subbands. Existing techniques rely on either a priori knowledgeof the audio processing to calculate the compensation filters ornon-real-time measurements of amplitude/phase characteristics of theprocessing in the transition band of the subbands.

SUMMARY

This Summary is provided to introduce a selection of concepts in asimplified form that are further described below in the DetailedDescription. This Summary is not intended to identify key features oressential features of the claimed subject matter, nor is it intended tobe used to limit the scope of the claimed subject matter.

Embodiments of the multi-rate arbitrary audio processing system andmethod described herein are an efficient high sampling rate reproductionsystem for audio processing designed for lower sampling rates. To reducecomputational complexity, high sampling rate signals are decimated andsplit into two subbands. The process may be repeated in the lowestsubband to obtain a maximally decimated system. Embodiments of thesystem and method apply audio processing only to the lowest frequencysubband to reduce computational complexity while preserving the fullbandwidth of the original content in the reconstruction phase. The audioprocessing may be arbitrary; known or unknown, linear, non-linear ortime-variant, or subject to user changes to various control parameters.Embodiments of the system and method are particularly well suited foraudio processing such as non-linear processing or user set controlparameters that produce time-varying or undetermined amplitude and phasechanges. Embodiments of the system and method are well suited forarbitrary linear audio processing as the embodiments can be universallyapplied without having to redesign filters for each audio processing.

To overcome the amplitude and phase changes caused by audio processingin the low band, embodiments of the multi-rate arbitrary audioprocessing system and method use real-time amplitude and phasemeasurement and compensation methods for arbitrary audio processing toimprove accuracy. In particular, amplitude and phase measurements areperformed in the transition band of the lowest frequency subband andcompensation is performed on the rest of the subbands to reduce aliasingnoise and amplitude distortion that can be caused when the subbands arerecombined. Embodiments of the audio processing process blocks of audiosamples. The measurement and compensation methods suitably process thesame blocks of audio samples to update the amplitude and phasecompensation for each block. The subbands are up-sampled and recombined.

Embodiments of the system and method provide for real-time measurementand compensation by determining whether there is sufficient signalamplitude and tonal content in the transition band of the lowestfrequency subband to reliably calculate the change in amplitude andphase caused by audio processing. If so, the amplitude and phasecompensation are calculated and applied to the rest of the bands. Ifnot, either the last calculated amplitude and phase compensation areapplied to the rest of the subbands or the compensation is disableduntil sufficient amplitude and tonal content is present in thetransition band.

Embodiments of the system and method provide for real-time measurementand compensation by determining whether there is sufficient signalamplitude and tonal content in the transition band of the lowest subbandto reliably calculate the change in amplitude and phase. If so, theamplitude and phase compensation are calculated and applied to the restof the subbands. If not, one or more tone signals positioned in thetransition band are added (injected) to the lowest subband before audioprocessing and subtracted (removed) from the lowest subband after audioprocessing. The one or more tone signals provide the tonal content andamplitude required to reliably calculate the amplitude and phasecompensation. Embodiments measure the signal amplitude and phase in thelowest subband, determine the amplitude of the tone signal(s) to behigher than a noise signal in the transition band but low enough to makeminimal change to the signal amplitude in the lowest subband and selectthe phase and frequency of the tone signal(s) to avoid cancellation ofthe signal in the transition band. Embodiments may remove signal in thetransition band prior to audio processing and inject the signal back inafter audio processing to provide clean measurements and calculationsbased only on the effects of audio processing on the one or more tonesignals.

Embodiments of the system and method inject multiple tone signals spacedat different frequencies in the transition band. Embodiments calculate aweighted average of the measurements for the multiple tone signals toproduce a single amplitude compensation value and a single delaycompensation value. Embodiments calculate one or more compensationfilters whose amplitude and phase characteristics approximately matchthe amplitude and phase characteristics of the audio processing in thetransition band. Amplitude/delay compensation is more computationallyefficient whereas the compensation filter provides more preciseamplitude and phase compensation. The compensation filter may berecalculated for each block of audio processed samples or lessfrequently at a defined interval.

Embodiments of the system and method compare amplitudes of the fulllowest frequency subband measurement (e.g. RMS) and the transition bandmeasurement to achieve balance between preservation of full band signallevel or intended gain of the audio processing (full subbandmeasurement) and perfect reconstruction characteristics (transition bandmeasurement).

These features allow embodiments of the system and method to operate atlow computational cost as compared to the cost of full high samplingrate audio processing, or to the cost of a QMF system with phasecompensation filtering. In addition, using the novel transition bandaddition/subtraction method reduces amplitude distortion and aliasingnoise in the reconstructed signal by a considerable amount.

It should be noted that alternative embodiments are possible, and stepsand elements discussed herein may be changed, added, or eliminated,depending on the particular embodiment. These alternative embodimentsinclude alternative steps and alternative elements that may be used, andstructural changes that may be made, without departing from the scope ofthe invention.

DRAWINGS DESCRIPTION

Referring now to the drawings in which like reference numbers representcorresponding parts throughout:

FIG. 1 is a block diagram of an embodiment of an M-channel QuadratureMirror Filter (QMF) Bank Audio processing structure with adecimation/expansion ratio of M configured for real-time measurement andcompensation of amplitude and phase changes caused by audio processingin the lowest frequency subband.

FIG. 2 is a block diagram of an embodiment of a 2-channel QMF Bank Audioprocessing structure in which real-time measurement and compensation isachieved by adding and then subtracting multi-tone signals into thelowest frequency subband before and after the audio processing.

FIG. 3 is a diagram illustrating the frequency response of the lower andupper frequency subbands in the 2-channel structure and the injection ofmulti-tone signals in the transition band that separates the lower andupper subbands.

FIG. 4 is a block diagram illustrating the analysis filter bank inpolyphase representation of a 2-channel QMF bank.

FIG. 5 is a block diagram illustrating the synthesis filter bank inpolyphase representation of a 2-channel QMF bank.

FIG. 6 is a diagram illustrating the real-time amplitude compensationbased on a comparison of amplitude measurements in the transition bandand across the full lowest frequency subband;

FIG. 7 is a flow diagram illustrating the decision process for injectingtone signals into the transition band and the calculation of theamplitude/phase of the tone signals.

FIG. 8 is a block diagram of an embodiment of a 2-channel QMF Bank Audioprocessing structure in which real-time measurement is used to update acompensation filter(s) for the higher frequency subbands.

DETAILED DESCRIPTION

In the following description of embodiments of a multi-rate arbitraryaudio processing system and method reference is made to the accompanyingdrawings. These drawings shown by way of illustration specific examplesof how embodiments of the multi-rate arbitrary audio processing systemand method may be practiced. It is understood that other embodiments maybe utilized and structural changes may be made without departing fromthe scope of the claimed subject matter.

The existing techniques for off-line design of compensation filters forthe high frequency subband(s) to correct the amplitude/phase changescaused by audio processing in the lowest frequency subband areunsuitable for a standalone system to be used with arbitrary audioprocessing. The compensation filters are designed and fixed based eitheron a priori knowledge of the audio processing or offline measurements ofthe induced amplitude/phase changes for each audio processinginstantiation. Audio processing such as non-linear processing, lineartime-variant processing or linear or non-linear processing having userset control parameters produce time-varying or undetermined amplitudeand phase changes. The existing techniques do not adequately addresssuch conditions. Furthermore, even with true linear audio processing, inwhich the amplitude and phase compensation should be fixed, existingtechniques require a specific filter design for each audio processingtechnique.

The present techniques that provide for real-time measurement of theamplitude/phase changes in the transition band of the low subband causedby the audio processing and application of amplitude/phase compensationto the higher subband(s) are well suited for a standalone system to beused with arbitrary audio processing. The real-time measurement andcompensation adapts to time-varying amplitude/phase changes regardlessof the source of the change (e.g. non-linear, time-varying linear oruser control parameters) and provides universal applicability for anylinear audio processing.

The system details of components of embodiments of the multi-ratearbitrary audio processing system and method will now be discussed. Itshould be noted that only a few of the several ways in which thecomponents, modules, and systems may be implemented are detailed below.Many variations are possible from that which is shown. For example, insome configurations the subband signals in each of the subbands aredownsampled/upsampled. In other configurations, the subband signal inthe lowest subband is downsampled to the sampling rate of the audioprocessing and then upsampled but the subband signals in the high orhigher subband may or may not be downsampled/sampled. Either approachcan be implemented in a QMF bank although a polyphase form of the QMFrequires all of the subbands to be downsampled.

Referring now to FIG. 1, an embodiment of an audio reproduction system10 includes a microphone 12 that converts sound 14 (pressure waves) intoan analog signal and an analog-to-digital (A/D) converter 16 thatsamples the analog signal to produce a high sampling rate digital audiosignal x(n) 17 (mono, stereo or multi-channel). Sampling rates of 88.2kHz, 96 kHz, 176.4 kHz and 192 kH being exemplary. Alternately, thedigital audio signal can be pre-recorded and stored as a digital audiofile 18. The high sampling rate digital audio signal is input to amulti-rate audio processing system 20 that implements subbanddecomposition, decimation, expansion and synthesis processes. Audioprocessing is applied to the downsampled audio signal in only the lowestsubband. Typical sampling rates being 44.1 kHz and 48 kHz.Amplitude/phase changes caused by the audio processing are measured inreal-time and compensation for such changes is applied to the highersubbands in real-time in a manner that is independent of the particularaudio processing performed (i.e. non-linear, linear time-variant,control parameters or a specific linear design). The multi-rate audioprocessing system 20 may be implemented with one or more IC chips 22,each configured with one or more digital processors 24 and memory 26.The reconstructed digital audio signal x′(n) is passed through adigital-to-analog (D/A) converter 28 to produce an analog signal thatdrives a loudspeaker 30 to produce sound 32 (pressure waves).Alternately, the digital audio signal x′(n) can be stored as a digitalaudio file 33.

Multi-rate audio processing system 20 is implemented as an M-channelQuadrature Mirror Filter (QMF) bank audio processing structure 34 with adecimation/expansion ratio of M, where M is an integer value. Analysisfilters H₀(z), H₁(z) H_(m-1)(z) 36 a, 36 b . . . 36 _(M-1) decomposeinput digital audio signal x(n) into M frequency subbands, which overlapin a transition band, to generate M subband signals. Decimators 38 a, 38b . . . 38 _(M-1) decimate the respective subband signals to generate Mdownsampled subband signals.

An audio processor 40 performs audio processing on the downsampledsubband signal in the lowest subband to generate a first processedsignal. Typically, the subband signal is segmented into blocks of audiosamples (e.g. 256, 512, 1024 samples) and each block is processed. Audioprocessing may constitute a linear process such as linearfiltering—lowpass, highpass and bandpass filtering, which may be fixedor time-varying, a non-linear process such as dynamic range compression,limiter and modulation, or user control parameters such as graphicequalizer gain, processing enable/disable and filter cutoff frequency.The audio processing produces amplitude and phase changes to thedownsampled audio signal in the transition band. The audio processingalso imparts an intended gain on the signal for the full lowestfrequency subband.

A compensation unit 42 measures in real-time amplitude and phase in thetransition frequency band of the downsampled subband signal (before andafter audio processing), processes the measurements to calculate achange in amplitude and phase in the transition frequency band andcalculates both an amplitude compensation and a phase compensation forthe remaining higher frequency subbands to approximately match themeasured change in amplitude and phase in the lowest frequency subband.The amplitude compensation may be calculated based only on the change inamplitude in the transition band, in which case the compensationprovides for perfect reconstruction of the output audio signal.Alternately, the amplitude compensation may be calculated based on thechange in amplitude in both the transition band and the entire lowestfrequency subband to balance perfect reconstruction of the output audiosignal and preservation of the intended gain or signal level of theaudio processing. The measurements and calculations are suitablyperformed in only the transition frequency band to enhance computationalefficiency.

The amplitude and phase compensation are applied to each of thedownsampled subband signals in the higher frequency subbands to generateprocessed subband signals. In an embodiment, the compensation isprovided in the form of one or more compensation filters 44. In anotherembodiment, compensation is provided in the form of a single value gain46 (amplitude compensation) and a single value delay 48 (phasecompensation. In a filter configuration, additional delay is provided tocompensate for processing latency of the audio processing. In thegain/delay configuration, the delay 48 includes both the group delay andthe processing latency. In an M-channel implementation, the delay ineach higher subband is an integer multiple of the calculated group delayplus the group delay of the filter banks. Amplitude/delay compensationis more computationally efficient whereas the compensation filterprovides more precise amplitude and phase compensation.

For each block of processed audio samples, compensation unit 42preferably determines whether there is sufficient amplitude and tonalcontent of the downsampled subband signal in the transition band of thelowest subband to reliably calculate the change in amplitude/phase,hence the amplitude/phase compensation. If the signal is too small ortoo noisy, the calculations are not valid. If this is the case,compensation unit 42 may be configured to implement various options. Inone option, compensation unit 42 simply disables the calculation andapplication of amplitude/phase compensation until a strong tonal signalin the transition band is detected. In another option, compensation unit42 disables the calculation of new amplitude/phase compensation andapplies the last calculated amplitude/phase compensation until a strongtonal signal is detected. In yet another option, the compensation unitcan insert one or more tone signals into the transition band of thedownsampled subband signal for the lowest subband to create a strongtonal signal for measurement purposes. The one or more tone signals arethen removed from the downsampled subband signal after audio processing.The tone signals are suitably selected to make minimal changes to thesignal amplitude in the lowest frequency subband and to avoid cancelingexisting signal in the transition band. To provide clean tonal signalsfor measurement and processing, the compensation unit may remove theaudio signal in the transition pre audio processing and re-insert thesignal after audio processing. In an embodiment, this may beaccomplished by performing a full FFT on the block of samples, removingthe signal in the transition band and performing an inverse FFT.

After processing, expanders 50 a, 50 b, . . . 50 _(m-1) expand theprocessed subband signals in the M channels by a factor of M. Synthesisfilters F₀(z), F₁(z) . . . F_(m-1)(z) process the respective subbandsignals, which are recombined 54 to generate a digital audio outputsignal x′(n), which is either saved as a digital audio file or convertedto sound.

Referring now to FIG. 2, an embodiment of a multi-rate audio processingsystem 98 is implemented as a two-channel Quadrature Mirror Filter BankAudio processing structure 100 with a decimation/expansion ratio of 2.Following two analysis filters, H₀(z) 110 and H₁(z) 115, the inputsignal x(n) 120 is decimated 130, 135 by a factor of 2. If the system ismaximally decimated the sampling rate of the subbands becomes the sameas the sampling rate of the audio processing 140 and the audioprocessing 140 can be performed only in the low band. Gain, phase andlatency compensations are required in the high band as the audioprocessing 140 in the low band can introduce amplitude/phase changes aswell as delay. In this embodiment, gain compensation 145 and delaycompensation 150 (phase (group delay) and latency) are applied to thehigh subband. The processed subband signals are expanded 160, 165 by afactor of 2, processed with synthesis filters F₀(z) 170 and F₁(z) 175,and recombined 180 to generate a digital audio signal x′(n) 185.

As shown in FIG. 3, the analysis filters H₀(z) 110 and H1(z) 115 andsynthesis filters F₀(z) 170 and F₁(z) 175 are lowpass and highpassfilters, respectively, with crossover frequency at π/2. The lowpassfilter has a pass band 187, a transition band 188 centered at thecrossover frequency and a stop band 189. The highpass filter has a stopband 190, a transition band 191 centered at the crossover frequency anda pass band 192.

Mathematically, the relationship between analysis filters H₀(z) 110 andH₁(z) 115 can be described as,H ₁(z)=H ₀(−z)  (1).

In order to cancel aliasing, the synthesis filters F₀(z) 170 and F₁(z)175 need to meet the following conditions:F ₀(z)=H ₁(−z),F(z)=−H ₀(−z)  (2).

Equations (1) and (2) indicate that aliasing free analysis and synthesisfilters can be designed from a single filter H₀(z) 110.

Representing Quadrature Mirror Filter (QMF) banks in polyphase formoffers computational benefits. Polyphase representation is a method thatseparates filter coefficients into multiple groups. For a ratio of 2,the even numbered filter coefficients are separated from the oddnumbered ones. Using polyphase representation, the analysis filter H₀(z)110 can be written as,H ₀(z)=E ₀(z ²)+z ⁻¹ E ₁(z ²)  (3).

Using equations (1), (2) and (3), equations (4) and (5) are obtained asfollows:

$\begin{matrix}{\begin{bmatrix}{H_{0}(z)} \\{H_{1}(z)}\end{bmatrix} = {\begin{bmatrix}1 & 1 \\1 & {- 1}\end{bmatrix}\begin{bmatrix}{E_{0}\left( z^{2} \right)} \\{z^{- 1}{E_{1}\left( z^{2} \right)}}\end{bmatrix}}} & (4) \\{\begin{bmatrix}{F_{0}(z)} \\{F_{1}(z)}\end{bmatrix} = {{\begin{bmatrix}1 & 1 \\1 & {- 1}\end{bmatrix}\begin{bmatrix}{z^{- 1}{E_{1}\left( z^{2} \right)}} \\{E_{0}\left( z^{2} \right)}\end{bmatrix}}.}} & (5)\end{matrix}$

If the decimation/expansion ratio is 2, E₀(z²) and E₁(z²) in equations(4) and (5) can be transposed using decimation and interpolation nobleidentities. They then become E₀(z) and E₁(z). This results in thecorresponding filters operating at the lower rate. FIG. 4 is a blockdiagram illustrating the analysis filter bank 110, 115 in polyphaserepresentation of the QMF banks. FIG. 5 is a block diagram illustratingthe synthesis filter bank 170, 175 in polyphase representation of theQMF banks.

The polyphase representation of QMF can be expanded to amulti-resolution structure to achieve a maximally decimated system. Thesignal is decimated and divided into two subbands, and the same processis applied to the low band signal. As audio processing is only performedin the lowest band, the high band of the first subband does not requiresubband processing. The terms, E₀(z) and E₁(z), in FIGS. 4 and 5 arereplaced with allpass filters A₀(z) and A₁(z) given the power symmetricfilter design described above. The terms A₀(z)/A₁(z) and A′₀(z)/A′₁(z)may not be the same filters subject to the design requirements of thesystem.

In addition to aliasing cancellation, it is also desirable to preventamplitude distortion when reconstructing the signal. If E₀(z) 200 andE₁(z) 205 in FIGS. 4 and 5 are Infinite Impulse Response (IIR) allpassfilters, this source of amplitude distortion is eliminated. The twoallpass filters for E₀(z) 200 and E₁(z) 205 are obtained by designingthe analysis filter H₀(z) 110 to be a power symmetric filter anddecomposing it into two allpass filters.

Power symmetric IIR halfband filters satisfy the following twoconditions:ω_(p)+ω_(s)=π  (6)δ_(p)=1−√{square root over (1−δ_(s) ²)}  (7)

Butterworth filters designed with cutoff frequency ω_(c)=0.5π satisfythe power symmetric conditions. Elliptic filters can also meet theconditions given in equations (6) and (7), but their parameters need tobe adjusted. The power symmetric elliptic filter design process isdescribed in detail in Chapter 7 (“Lth-band digital filters”) of thebook “Multirate Filtering for Digital Signal Processing MATLABApplications” by Ljiljana Milic, New York, Information ScienceReference, pages 206-237 (2009). Because the poles of power symmetricelliptic filters are located on the imaginary axis of the complex plane,the allpass filters for E₀(z) 200 and E₁(z) 205 are obtained using thepole interlacing property, which is described in Chapter 5 (“MaximallyDecimated Filter Banks”) of the book “Multirate Systems and FilterBanks” by P. P. Vaidyanathan, New Jersey, PTR Prentice-Hall, Inc., pages188-256 (1993).

Embodiments of the multi-rate arbitrary audio processing system andmethod are free of aliasing noise and amplitude distortion in partbecause they include a QMF system design with power symmetric filters.However, aliasing noise and amplitude distortion can be introduced as aresult of applying audio processing to the low subband. Even withamplitude and phase compensations made to the higher subbands tocompensate for the changes induced by low band processing it may not bepossible to eliminate amplitude distortion and aliasing noisecompletely. Therefore, it is desirable to have a narrow transition bandwith high stopband attenuation in order to minimize the region thatcontains amplitude distortion and aliasing noise.

In some embodiments of multi-rate arbitrary audio processing system andmethod the design criteria for the analysis filter H₀(z) 110 in a 2:1ratio decimation/expansion system are: (a) a stopband attenuation of 96dB or higher; and (b) a transition bandwidth of 0.4167π to 0.5833π.

The passband edge frequency 0.4167π is equivalent to 20 kHz at 48 kHzsampling rate. Therefore, if aliasing noise and amplitude distortionexist, they become present in the frequency range above human hearing.As analysis filters H₀(z) 110 and H₁(z) 115 are a mirror image of eachother, amplitude distortion and aliasing noise will be less than −96dBFs with the stopband attenuation criterion set above. Embodiments ofthe multi-rate arbitrary audio processing system and method include anoptimal power symmetric filter that meets the filter design criteria.

In different embodiments, 13th order half band Butterworth and ellipticfilters have a passband edge at 0.4167π. The Butterworth filterattenuation is higher than that of the elliptic filter above 0.78πbecause of its monotonicity in the stopband. However, the Butterworthfilter involves a filter order increase in order to reduce thetransition bandwidth. Therefore, elliptic filters are more suitable andwere used in embodiments of the multi-rate arbitrary audio processingsystem and method given their lower filter order requirement for thetransition bandwidth criterion described above.

Given the choice of elliptic filter design some tradeoffs need to beconsidered. These are tradeoffs between transition bandwidth andstopband attenuation, and filter order and ripple size respectively. Asmaller transition band results in lower stopband attenuation.Increasing the filter order can produce higher stopband attenuation, butit will increase the ripple size and also the computational cost. Theoptimal design for the given filter design criteria is a 13^(th) orderelliptic filter with passband edge 0.42π. Using the allpassdecomposition described above, E₀(z) and E₁(z) in FIGS. 4 and 5 are3^(rd) order allpass filters for a 13^(th) order elliptic filter.

Referring again to FIG. 2, in order to compensate the higher subbandsfor the amplitude, phase and delay caused by audio processing in thelowest subband, real-time measurements of the amplitude and phase of thesignal in the lowest frequency subband must be made and processed todetermine whether the audio signal itself has sufficient tonal contentand signal strength in the transition band to reliably determine thecompensation and, if not what tone signals to inject, calculate thechange in amplitude in phase caused by audio processing and to calculatethe amplitude and phase compensation for the higher subbands in the formof either compensation filter(s) plus delay (processing latency) or asingle gain value and a single delay value (group delay plus processinglatency). Amplitude and phase compensation is not required in the fullbandwidth of the signal, only in the transition band. A compensationprocessor(s) 300 is configured to implement the measurement, control andcalculation techniques to provide in real-time the gain and phasecompensation. In an embodiment, processor 300 processes the each blockof samples (processed by the audio processing) to provide updated gainand phase compensation for each block.

Compensation processor(s) 300 are suitably configured to make real-timemeasurements of amplitude and phase of the signal in the lowestfrequency subband (full band and transition band) pre- andpost-injection of tone signals 302 and 304 and post audio processing306. The full band amplitude can be calculated as the root-mean-square(RMS) of the audio signal samples x(n) in the lowest frequency subbandfor each processed block of samples. The transition band amplitude canbe calculated as the average of Discrete Fourier Transform coefficientsover the transition band. To improve computational efficiency a Goertzelalgorithm can be used to compute the DFT coefficients. The Goertzelalgorithm has the form of a digital filter that provides the DFT valueof a given frequency. The phase of the signal can be calculated bytaking the inverse tangent of the imaginary part of the DFT term dividedby the real part.

Compensation processor(s) 300 are configured to process the full andtransition band amplitude and phase measurements to determine whether toinject one or more tone signals into the transition band of the audiosignal in the lowest frequency subband and, if so, the proper amplitude,phase and frequency of the tone signals 308. If the signal hassufficient amplitude and tonal content in the transition band, injectionof tone signals is not necessary. If tone signals are to be injected,the processor 300 sets the amplitude of the one or more tone signals tobe higher than a noise signal in the transition band but low enough tomake minimal change to the amplitude in the full first frequency subbandand sets the phase and frequency of the one or more tone signals toavoid cancellation of the first downsampled subband signal in thetransition band. The processor(s) are configured to implement amulti-tone signal generator 310 that injects the one or more tonesignals at a summing node 312 to add the tone signals to the audiosignal in the lowest frequency subband.

These tone signals are suitably single frequency discrete sine waves.Mathematically, the signal added to the low band before processing, andthe cancellation signal intended for subtracting the signal added aregiven by

$\begin{matrix}{{a(n)} = {\beta{\sum\limits_{k = \rho}^{N/2}{\cos\left( {2\;\pi\;{n\left( \frac{k}{N} \right)}} \right)}}}} & \left( 8 \right. \\{{c(n)} = {{- \beta}{\sum\limits_{k = \rho}^{N/2}{{\delta_{mag}(k)}{\cos\left( {{2\;\pi\;{n\left( \frac{k}{N} \right)}} + {\delta_{phase}(k)}} \right)}}}}} & (9)\end{matrix}$where, ρ is set to N/2−1 as the real-time amplitude and phasecompensation implementation only involves the group delay measurement atthe Nyquist frequency. Still, ρ can be set to the DFT point of thepassband edge frequency of the low band filter H₀(z) in order to exploitthe full transition band amplitude/phase information. The signalamplitude β is determined by the first measurement of the low bandsignal before the audio processing 140. The terms δ_(mag) (k) andδ_(phase)(k) are magnitude and phase differences calculated from themeasurements before and after the audio processing. As depicted in FIG.2, gain 314 and delay 316 are applied to the tone signals, which areremoved from the audio signal in the lowest frequency subband at summingnode 318. The signal added for the measurement may not be cancelled outcompletely since there can be amplitude/phase measurements error andround off error introduced by the system.

Compensation processor(s) 300 are configured to calculate the change inamplitude and phase in both the full band and transition band caused bythe audio processing 320. The change in the amplitude may be expressedas a ratio of the post-processing amplitude to the pre-processedamplitude. The change in phase may be expressed as the differencebetween the post and pre-processed phase measurements. The change inamplitude and phase is computed for each tone signal or at one or morefrequencies across the transition band (if tones are not injected)

Compensation processor(s) 300 are configured to calculate the amplitudeand phase compensation for the higher subbands to approximately matchthe measured change in amplitude and phase 322. As shown, theprocessor(s) calculate a single gain value to set gain 145 and a singledelay value (both group delay and processing latency) to set delay 150to apply the compensation to the signals in the higher frequencysubbands. Alternately, the processors may calculate a full compensationfilter(s) whose frequency response approximately matches the frequencyresponse in the transition band of the audio processing.

Applying root mean square (RMS) value changes of the low-band signal tothe high-band signal may satisfy the amplitude compensation requirementin terms of maintaining constant full band signal level, but it may notaccomplish perfect reconstruction of the signal in the QMF system ofembodiments of the multi-rate arbitrary audio processing system andmethod. As amplitude information of the signal in the transition band ofthe QMF system is obtained from the Goertzel algorithm, it can be usedto calculate amplitude changes in the transition band of the low bandsignal.

Equations (10) and (11) below show mathematically the comparison of RMSvalue changes and transition band level changes in order to achieve areasonable balance between preservation of full band signal level andperfect reconstruction characteristics.y _(h)(n)=m _(coef) x _(h)(n)  (10)m _(coef) =f(m _(δtr) ,m _(δRMS)),f(m _(δtr) ,m _(δRMS))=G(d _(δrms) _(_) _(tr),τ),  (11)d _(δrms) _(_) _(tr) =m _(δRMS)(dB)−m _(δtr)(dB)where m_(coef) is the actual amplitude compensation value applied to thehigh band of the QMF system. f(m_(δtr), m_(δRMS)) is a function thatcalculates the amplitude compensation value based on the transition bandamplitude change (m_(δtr)) and the lowest frequency subband change(m_(δRMS)). This function f( ) can be interpreted as function G( ) thattakes inputs of the amplitude change difference −d_(δrms) _(_) _(tr) anda target threshold to determine the balance between the transition bandand the first frequency subband. Function G( ) can be designed in manyways to address particular concerns for given applications. Equation(12) is an example of function G( )

$\begin{matrix}{{G\left( {d_{\delta{rms}\_{tr}},\tau} \right)} = \begin{Bmatrix}{m_{\delta{tr}},{d_{\delta{rms}\_{tr}} < \tau_{1}}} \\{{\left( {m_{\delta{tr}} + m_{\delta{RMS}}} \right)/2},{\tau_{2} > d_{\delta{rms}\_{tr}} > \tau_{1}}} \\{m_{\delta{RMS}},{d_{\delta{rms}\_{tr}} > \tau_{2}}}\end{Bmatrix}} & (12)\end{matrix}$In this example, the threshold τ is a set of two thresholds τ₁ and τ₂.In addition to the threshold settings, a smoothing algorithm is appliedto m_(coef) in a real-time implementation in order to ensure smoothamplitude transition over time.

Referring now to FIG. 6, an input signal 400 is split into lower andupper frequency subbands 402 and 404, respectively. Audio processing isperformed on the signal in the lower frequency subband to produce aprocessed subband signal 406. In this example, compensation is providedby a filter, hence may vary across the transition band. Amplitudecompensation is applied to signal 404 in the higher frequency subband inone of three ways. First, amplitude compensation 408 is computed basedonly on the amplitude change in the transition band and applied tosignal 404 to produce compensated signal 409. As shown in thecorresponding synthesized output signal 410, this approach providesperfect reconstruction as shown in 420, but does not maintain the fullband signal level (gain of audio processing) shown in 406. Second,amplitude compensation 412 is computed based only on the RMS value forthe full band and applied to signal 404 to produce compensated signal413. As shown in the corresponding synthesized output signal 414, thisapproach maintains the full band signal level (gain of audio processing)as shown in 406 but does not provide perfect reconstruction. Lastly,amplitude compensation 416 is computed using equation 11 to balanceperfect reconstruction and full band signal level and applied to signal404 to produce compensated signal 417. This balance is achieved as shownin the synthesized output signal 418.

As magnitude and phase response of the transition band for compensationare known from measurements and calculation, one can design a filter orfilters given magnitude and phase response. Alternatively, one candesign separate filters for magnitude matching and phase matching.Number of tones added to the transition band determines the frequencyresolution of the transition band. When designing compensation filter(s)the magnitude and phase response may need interpolations if thefrequency resolution is lower than it is required for the filter design.Often, the term phase compensation is considered to be the same as thegroup delay compensation. However, group delay can be measured not justusing adjacent frequency bins but frequency bins with bigger intervals.Therefore, group delay can be an approximation of phase change givenfrequency resolution.

While there are various methods available for group delay measurementsof discrete time signals, embodiments of the multi-rate arbitrary audioprocessing system and method use the direct differentiation method. Thedirect differentiation method calculates the derivative of the unwrappedphase of the signal. Mathematically this can be written as:

$\begin{matrix}{\tau_{g} = {- {\frac{d\;{\phi(\omega)}}{d\;\omega}.}}} & (13)\end{matrix}$

Phase changes caused by arbitrary audio processing in the low band ofthe QMF are measured using the Goertzel algorithm. The group delay atthe Nyquist frequency is then calculated and rounded to the nearestinteger value due to its greater amount of squared-magnitude overlapbetween the low and high bands in the QMF system. If arbitrary audioprocessing in the lowest subband of the QMF system introduces latency,the signal in the higher subbands must also be delayed to match thelatency introduced in the lowest subband. Since the delay block needs tobe there for latency compensation, the integer number group delaycompensation method adds no additional computation but a small amount ofmemory.

With one pure sine wave tone, we cannot do group delay matching as itrequires two DFT components to calculate group delay. But with one tonewe can do phase matching at Nyquist frequency as Nyquist frequency phasechange will always appear to be 0 degrees or 180 degrees.=>0 degrees: nocompensation, 180 degrees: 1-sample delay to make it 0 degrees.

Considering that a 1-sample delay at the Nyquist frequency is equivalentto a 180-degree phase shift for discrete time signals, an odd numbergroup delay will cause a 180-degree phase shift at the Nyquist frequencyin the high band of the QMF system. As power symmetric elliptic filtershave a 90-degree phase difference between the low band and the high bandat the Nyquist frequency, adding an odd number group delay does notcancel out the signals at Nyquist but changes the polarity of thesignals in the reconstruction phase.

However, the phase difference between the aliasing noise of the QMFsystem in the low band and that in the high band is 180 degrees. Assuch, the aliasing noise is cancelled out in the reconstruction phase.Therefore, adding an odd number group delay in the high band canactually increase the aliasing noise and amplitude distortion.Consequently, it is important to check the phase response change atNyquist.

For example, for a 0-degree phase shift and an odd number group delay atNyquist frequency, the group delay value should be rounded to thenearest even number integer value instead of the nearest integer numberif the Nyquist frequency has a 0-degree phase shift and odd number groupdelay.

Referring now to FIG. 7, an embodiment of step 308, tone injectiondetermination, in FIG. 2 first determines whether or not to inject tonesignals in the transition band of the lowest frequency subband and, ifyes, the amplitude and phase of the injected tones. The compensationprocessor is configured to determine whether the downsampled signal inthe transition band of the lowest frequency subband has both sufficientamplitude and sufficient tonal content to compute the amplitude andphase compensation. If not, the compensation processor is configured todetermine the amplitude of the tone signals to be higher than a noisesignal in the transition band but low enough to make minimal change tothe amplitude in the full lowest frequency subband. The compensationprocessor is configured to select the phase and frequency of the tonessignals to avoid cancellation of the downsampled signal in thetransition band.

In an embodiment, the compensation processor is configured to implementthree processes: Process A, Process B and Process C to determine whetherto inject tone signals and if so the amplitude and phase of the tonesignals. The compensation processor is configured to receive transitionband amplitudes (Amp[ ]) 500 and transition band phases (Phs[ ]) 502 forthe current block of processed samples.

The compensation processor is configured to process the transition bandamplitudes to determine whether the amplitude in the transition band isgreater than a minimum 504 (Process A). An example of Process Acomputes: Minimum amplitude(a)>(−6*B_(sys))(dB)+E_(md) (dB) whereB_(sys)=system bit resolution (ex—32 bit, 24 bit, 16 bit etc.) andE_(rnd)=Round-off error from calculation and Minimum amplitude (a) isthe minimum amplitude in transition band. In this example, the minimumamplitude required to calculate the amplitude/phase compensation is thesum of the minimum signal value for system bit resolution in dB and theexpected round-off error from the amplitude/phase calculation.

The compensation processor is configured to process the transition bandamplitudes to determine whether the transition band is tonal 506(Process B). An example of Process B computes spectral flatness of thetransition band to determine whether the magnitude spectrum is likely tocontain tonal components. The process measures the phase variation ofeach frequency in the transition band to measure the phase continuitiesthat can distinguish noise signal from tonal signal. By combiningspectral flatness and phase variation one can determine if the signalcontains tonal signal at a given frequency bin within the transitionband. For example,

$\begin{matrix}{{{{{Tonal}(k)} = {True}},{{{{Delta\_ p}(k)} < {Treshold\_ p}}\&}}\mspace{14mu}} \\{{{Spectral}\mspace{14mu}{flatness}\;(k)} > {Trehosld\_ sf}} \\{{= {False}},{{{Delta\_ p}(k)} > {{Treshold\_ p}\mspace{11mu}{\;}}}} \\{{{Spectral}\mspace{14mu}{{flatness}(k)}} < {Trehosld\_ sf}}\end{matrix}$ Treshold_p = phase  variation  thresholdTreshold_s = spectral  flatness   threshold

-   Spectral flatness(k)=Pl(k)/Avr, where k=1 . . . m, m=number of    peaks, calculated using Amp[ ]-   Pl(1, 2, . . . m)=Levels of spectral peaks in the transition band,    (m<n, m=number of peaks in the transition band, n=number of    transition band frequency bins)-   Avr=average level of the transition band amplitude.-   Delta_p(k)=Delta_p1(k)−Delta_p0(k): phase variation (k=1, 2, . . .    m), calculated using Phs[ ]-   Delta_p0(k)=abs(Phs[k]−Phs[k−1]), and-   Delta_p1(k)=abs(Phs[k+1]−Phs[k]).

If the transition band signal has both sufficient amplitude and tonalcontent (i.e. Process A and Process B are both “true”) 508, thecompensation processor does not inject tone signals 510 and processesthe transition band signal to determine the amplitude and phasecompensation. If the transition band signal lacks either sufficientamplitude or tonal content (i.e., Process A or Process B is “false) 508,the compensation processor is configured to process the transition bandamplitudes, transition band phases and the full lowest frequency subbandamplitudes 510 to calculate the amplitude/phase of the tones to beinjected 512 and inject the tones into the transition band 514.

In an example, the compensation processor is configured to set theminimum change of signal level, and calculate a minimum level of tone(s)using the full band amplitude of the input signal. For example

Find a minimum x_(rms) meets the requirement of 20log₁₀(In_(rms)+x_(rms)/In_(rms))<τ(dB)

Where In_(rms)=measured value of the lowest frequency subband amplitudeand x_(rms)=the level of the tone to be injected.

-   Case1 (a=true, b=false): amplitude is bigger but noise    -   τ=tone_noise_ratio: tone signal to noise ratio-   Case2 (a=false, b=true): amplitude is smaller but tonal    -   τ=min_level: the minimum level change threshold in Case2

${Case}\; 3\mspace{14mu}\left( {{a = {false}},{b = {false}}} \right)\text{:}\mspace{14mu}{amplitude}\mspace{14mu}{is}\mspace{14mu}{smaller}{\quad\mspace{14mu}{{{{and}\mspace{14mu}{noise}\text{}{calculate}\mspace{14mu} X_{rms}{for}\mspace{14mu}{both}\mspace{14mu}\tau} = {{{‘{min\_ level}’}\mspace{14mu}{and}\tau} = {{‘{{tone\_ noise}{\_ ratio}}’}\mspace{14mu}{and}}}},\begin{matrix}{{X_{{rm}s} = {X_{rms}\mspace{14mu}({min\_ level}\;)}},{{{if}\mspace{14mu} X_{rms}\mspace{14mu}({min\_ level}\;)} >}} \\{X_{rms}\mspace{14mu}\left( {{tone\_ noise}{\_ ratio}} \right)} \\{{= {X_{rms}\mspace{14mu}\left( {{tone\_ noise}{\_ ratio}} \right)}},{{{if}\mspace{14mu} X_{rms}\mspace{14mu}({min\_ level}\;)} <}} \\{X_{rms}\mspace{14mu}{\left( {{tone\_ noise}{\_ ratio}} \right).}}\end{matrix}}}$

Referring now to FIG. 8, an embodiment of a multi-rate audio processingsystem 600 is implemented as a two-channel Quadrature Mirror Filter Bankaudio processing structure with a decimation/expansion ratio of 2. Inthis embodiment, the amplitude/phase compensation in the higherfrequency subband is performed using one or more compensation filters.Compensation filters provide better matching of the amplitude and phasechanges induced in the transition band by the audio processing at thecost of additional computations.

Following two analysis filters, H₀(z) 606 and H₁(z) 608, the inputsignal x(n) 610 is decimated 612 and 614 by a factor of 2. Audioprocessing 616 is performed on a block of samples of the downsampledinput signal x(n). The amplitude/phase of the downsampled signal x(n)are measured before and after audio processing and processed 618 tocalculate a change in the amplitude and phase of the signal in thetransition band caused by the audio processing to provide fulltransition band frequency/phase response information. One or morecompensation filters are designed 620 from this information to provide afilter response that approximately matches the frequency/phase responsein the transition band. FIR or IIR filters can be used for designingfilters given magnitude (amplitude) and phase characteristics. It can bea same filter set that approximately matches the amplitude and phaseresponse, or separate filter sets for the amplitude matching and thephase matching. These designs are used to update the filters 622. Datasmoothing/interpolation 624 is applied to the updated filters to preventany artifacts that can be caused by updating filters or filtercoefficients. These interpolated/smoothed filters are applied to thestate variables of the filter processing in the highest frequencysubband to provide amplitude and phase compensation 626. Delay 628provides compensation for the processing latency of the audioprocessing. The processed subband signals are expanded 630 and 632 by afactor of 2, processed with synthesis filters F₀(z) 634 and F₁(z) 636,and recombined 638 to generate a digital audio signal x′(n) 640.

Many other variations than those described herein will be apparent fromthis document. For example, depending on the embodiment, certain acts,events, or functions of any of the methods and algorithms describedherein can be performed in a different sequence, can be added, merged,or left out altogether (such that not all described acts or events arenecessary for the practice of the methods and algorithms). Moreover, incertain embodiments, acts or events can be performed concurrently, suchas through multi-threaded processing, interrupt processing, or multipleprocessors or processor cores or on other parallel architectures, ratherthan sequentially. In addition, different tasks or processes can beperformed by different machines and computing systems that can functiontogether.

The various illustrative logical blocks, modules, methods, and algorithmprocesses and sequences described in connection with the embodimentsdisclosed herein can be implemented as electronic hardware, computersoftware, or combinations of both. To clearly illustrate thisinterchangeability of hardware and software, various illustrativecomponents, blocks, modules, and process actions have been describedabove generally in terms of their functionality. Whether suchfunctionality is implemented as hardware or software depends upon theparticular application and design constraints imposed on the overallsystem. The described functionality can be implemented in varying waysfor each particular application, but such implementation decisionsshould not be interpreted as causing a departure from the scope of thisdocument.

The various illustrative logical blocks and modules described inconnection with the embodiments disclosed herein can be implemented orperformed by a machine, such as a general purpose processor, aprocessing device, a computing device having one or more processingdevices, a digital signal processor (DSP), an application specificintegrated circuit (ASIC), a field programmable gate array (FPGA) orother programmable logic device, discrete gate or transistor logic,discrete hardware components, or any combination thereof designed toperform the functions described herein. A general purpose processor andprocessing device can be a microprocessor, but in the alternative, theprocessor can be a controller, microcontroller, or state machine,combinations of the same, or the like. A processor can also beimplemented as a combination of computing devices, such as a combinationof a DSP and a microprocessor, a plurality of microprocessors, one ormore microprocessors in conjunction with a DSP core, or any other suchconfiguration.

Embodiments of the multi-rate arbitrary audio processing system andmethod described herein are operational within numerous types of generalpurpose or special purpose computing system environments orconfigurations. In general, a computing environment can include any typeof computer system, including, but not limited to, a computer systembased on one or more microprocessors, a mainframe computer, a digitalsignal processor, a portable computing device, a personal organizer, adevice controller, a computational engine within an appliance, a mobilephone, a desktop computer, a mobile computer, a tablet computer, asmartphone, and appliances with an embedded computer, to name a few.

Such computing devices can be typically be found in devices having atleast some minimum computational capability, including, but not limitedto, personal computers, server computers, hand-held computing devices,laptop or mobile computers, communications devices such as cell phonesand PDA's, multiprocessor systems, microprocessor-based systems, set topboxes, programmable consumer electronics, network PCs, minicomputers,mainframe computers, audio or video media players, and so forth. In someembodiments the computing devices will include one or more processors.Each processor may be a specialized microprocessor, such as a digitalsignal processor (DSP), a very long instruction word (VLIW), or othermicrocontroller, or can be conventional central processing units (CPUs)having one or more processing cores, including specialized graphicsprocessing unit (GPU)-based cores in a multi-core CPU.

The process actions of a method, process, or algorithm described inconnection with the embodiments disclosed herein can be embodieddirectly in hardware, in a software module executed by a processor, orin any combination of the two. The software module can be contained incomputer-readable media that can be accessed by a computing device. Thecomputer-readable media includes both volatile and nonvolatile mediathat is either removable, non-removable, or some combination thereof.The computer-readable media is used to store information such ascomputer-readable or computer-executable instructions, data structures,program modules, or other data. By way of example, and not limitation,computer readable media may comprise computer storage media andcommunication media.

Computer storage media includes, but is not limited to, computer ormachine readable media or storage devices such as Blu-ray discs (BD),digital versatile discs (DVDs), compact discs (CDs), floppy disks, tapedrives, hard drives, optical drives, solid state memory devices, RAMmemory, ROM memory, EPROM memory, EEPROM memory, flash memory or othermemory technology, magnetic cassettes, magnetic tapes, magnetic diskstorage, or other magnetic storage devices, or any other device whichcan be used to store the desired information and which can be accessedby one or more computing devices.

A software module can reside in the RAM memory, flash memory, ROMmemory, EPROM memory, EEPROM memory, registers, hard disk, a removabledisk, a CD-ROM, or any other form of non-transitory computer-readablestorage medium, media, or physical computer storage known in the art. Anexemplary storage medium can be coupled to the processor such that theprocessor can read information from, and write information to, thestorage medium. In the alternative, the storage medium can be integralto the processor. The processor and the storage medium can reside in anapplication specific integrated circuit (ASIC). The ASIC can reside in auser terminal. Alternatively, the processor and the storage medium canreside as discrete components in a user terminal.

The phrase “non-transitory” as used in this document means “enduring orlong-lived”. The phrase “non-transitory computer-readable media”includes any and all computer-readable media, with the sole exception ofa transitory, propagating signal. This includes, by way of example andnot limitation, non-transitory computer-readable media such as registermemory, processor cache and random-access memory (RAM).

The phrase “audio signal” is a signal that is representative of aphysical sound.

Retention of information such as computer-readable orcomputer-executable instructions, data structures, program modules, andso forth, can also be accomplished by using a variety of thecommunication media to encode one or more modulated data signals,electromagnetic waves (such as carrier waves), or other transportmechanisms or communications protocols, and includes any wired orwireless information delivery mechanism. In general, these communicationmedia refer to a signal that has one or more of its characteristics setor changed in such a manner as to encode information or instructions inthe signal. For example, communication media includes wired media suchas a wired network or direct-wired connection carrying one or moremodulated data signals, and wireless media such as acoustic, radiofrequency (RF), infrared, laser, and other wireless media fortransmitting, receiving, or both, one or more modulated data signals orelectromagnetic waves. Combinations of the any of the above should alsobe included within the scope of communication media.

Further, one or any combination of software, programs, computer programproducts that embody some or all of the various embodiments of themulti-rate arbitrary audio processing system and method describedherein, or portions thereof, may be stored, received, transmitted, orread from any desired combination of computer or machine readable mediaor storage devices and communication media in the form of computerexecutable instructions or other data structures.

Embodiments of the multi-rate arbitrary audio processing system andmethod described herein may be further described in the general contextof computer-executable instructions, such as program modules, beingexecuted by a computing device. Generally, program modules includeroutines, programs, objects, components, data structures, and so forth,which perform particular tasks or implement particular abstract datatypes. The embodiments described herein may also be practiced indistributed computing environments where tasks are performed by one ormore remote processing devices, or within a cloud of one or moredevices, that are linked through one or more communications networks. Ina distributed computing environment, program modules may be located inboth local and remote computer storage media including media storagedevices. Still further, the aforementioned instructions may beimplemented, in part or in whole, as hardware logic circuits, which mayor may not include a processor.

Conditional language used herein, such as, among others, “can,” “might,”“may,” “e.g.,” and the like, unless specifically stated otherwise, orotherwise understood within the context as used, is generally intendedto convey that certain embodiments include, while other embodiments donot include, certain features, elements and/or states. Thus, suchconditional language is not generally intended to imply that features,elements and/or states are in any way required for one or moreembodiments or that one or more embodiments necessarily include logicfor deciding, with or without author input or prompting, whether thesefeatures, elements and/or states are included or are to be performed inany particular embodiment. The terms “comprising,” “including,”“having,” and the like are synonymous and are used inclusively, in anopen-ended fashion, and do not exclude additional elements, features,acts, operations, and so forth. Also, the term “or” is used in itsinclusive sense (and not in its exclusive sense) so that when used, forexample, to connect a list of elements, the term “or” means one, some,or all of the elements in the list.

While the above detailed description has shown, described, and pointedout novel features as applied to various embodiments, it will beunderstood that various omissions, substitutions, and changes in theform and details of the devices or algorithms illustrated can be madewithout departing from the spirit of the disclosure. As will berecognized, certain embodiments of the inventions described herein canbe embodied within a form that does not provide all of the features andbenefits set forth herein, as some features can be used or practicedseparately from others.

Moreover, although the subject matter has been described in languagespecific to structural features and methodological acts, it is to beunderstood that the subject matter defined in the appended claims is notnecessarily limited to the specific features or acts described above.Rather, the specific features and acts described above are disclosed asexample forms of implementing the claims.

What is claimed is:
 1. A method of audio reproduction, comprising:splitting a first digital audio signal into at least a first and asecond frequency subbands to generate at least a first and a secondsubband signal, the first and second frequency subbands separated by atransition frequency band; generating a first processed subband signalby performing an audio processing on the first subband signal; measuringamplitude and phase in the transition frequency band of the firstsubband signal and the first processed subband signal; processing themeasurements to calculate a change in amplitude and phase in thetransition frequency band; calculating both an amplitude compensationand a phase compensation for the second frequency subband toapproximately match the measured change in amplitude and phase in thefirst frequency subband; generating a second processed subband signal byapplying the amplitude and phase compensation to the transition band ofthe second subband signal; and combining the first and second processedsubband signals to reconstruct an output audio signal.
 2. The method ofclaim 1, wherein the amplitude and phase in the transition frequencyband of the first subband signal and the first processed subband signalare measured in real time.
 3. The method of claim 1, wherein theamplitude and phase in the transition frequency band of the firstsubband signal and the first processed subband signal are measuredduring initialization.
 4. The method of claim 1, wherein both theamplitude compensation and the phase compensation for the secondfrequency subband are calculated in real time.
 5. The method of claim 1,further comprising wherein both the amplitude compensation and the phasecompensation for the second frequency subband are calculated duringinitialization.
 6. The method of claim 1, wherein the second processedsubband signal is generated in real time.
 7. The method of claim 1,wherein the second processed subband signal is generated duringinitialization.
 8. A method of audio reproduction, comprising: splittinga first digital audio signal into at least a first and a secondfrequency subbands to generate at least a first and a second subbandsignal, the first and second frequency subbands separated by atransition frequency band; generating a first downsampled subband signalby downsampling or decimating the first subband signal; generating afirst processed subband signal by performing an audio processing on thefirst downsampled subband signal; measuring an amplitude and a phase inthe transition frequency band of the first downsampled subband signaland the first processed subband signal; processing the measurements tocalculate a change in amplitude and phase in the transition frequencyband; calculating both an amplitude compensation and a phasecompensation for the second frequency subband to approximately match themeasured change in amplitude and phase in the first frequency subband;generating a second processed subband signal by applying the amplitudeand phase compensation to the transition band of the second subbandsignal; and combining the first and second processed subband signals toreconstruct an output audio signal.
 9. The method of claim 8, whereinthe amplitude and the phase in the transition frequency band of thefirst downsampled subband signal and the first processed subband signalare measured in real time.
 10. The method of claim 8, wherein theamplitude and the phase in the transition frequency band of the firstdownsampled subband signal and the first processed subband signal aremeasured during initialization.
 11. The method of claim 8, wherein boththe amplitude compensation and the phase compensation for the secondfrequency subband are calculated in real time.
 12. The method of claim8, wherein both the amplitude compensation and the phase compensationfor the second frequency are calculated during initialization.
 13. Themethod of claim 8, wherein the second processed subband signal isgenerated in real time.
 14. The method of claim 8, wherein the secondprocessed subband signal is generated during initialization.
 15. Anaudio reproduction system, comprising: a quadrature mirror filter (QMF)bank configured to split a digital audio signal into at least a lowestand a highest frequency subbands to generate at least a first and asecond subband signal, the lowest and highest frequency subbandsseparated by a transition frequency band, and to decimate at least thefirst subband signal to generate a first downsampled subband signal; anaudio processor configured to perform audio processing on the firstdownsampled subband signal to generate a first processed subband signal;amplitude and phase compensators configured to apply amplitude and phasecompensation to the second subband signal to generate a second processedsubband signal; and a compensation unit configured to measure amplitudeand phase in the transition frequency band of the first downsampledsubband signal and the first processed subband signal before and afterthe audio processor, respectively, calculate a change in amplitude andphase in the transition frequency band, and calculate updates for theamplitude and phase compensation to approximately match the calculatedchange, wherein the QMF filter is configured to expand at least thefirst processed subband signal and to combine the first and secondprocessed subband signals to reconstruct an audio output.
 16. The audioreproduction system of claim 15, wherein the compensation unit isconfigured to measure in real time the amplitude and the phase inreal-time and calculate the updates for the amplitude and the phasecompensation in real time.
 17. The audio reproduction system of claim15, wherein the compensation unit is configured to measure the amplitudeand the phase during initialization and calculate the updates for theamplitude and the phase compensation during initialization.
 18. Theaudio reproduction system of claim 15, wherein the compensation unit isconfigured to measure in real time the amplitude and the phase inreal-time and calculate the updates for the amplitude and the phasecompensation in real-.
 19. The audio reproduction system of claim 15,wherein the compensation unit is configured to measure duringinitialization the amplitude and the phase in the transition frequencyband of the first downsampled subband signal and the first processedsubband signal before and after the audio processor, respectively,calculate the change in amplitude and the phase in the transitionfrequency band, and calculate in real time the updates for the amplitudeand the phase compensation to approximately match the calculated change.